1. Field of the Invention
The present invention relates to a switching power supply circuit equipped to various types of equipment as a power supply source.
2. Description of the Related Art
The applicant of this application previously proposed a technique of constructing a composite resonance type converter as a switching power supply circuit by combining a voltage resonance type converter of one stone at a primary side and a half-wave rectifying type voltage resonance circuit at a secondary side, providing an active clamp circuit at the secondary side and stabilizing the DC output voltage thereof by controlling the conduction angle of the switching element of the active clamp circuit.
FIG. 10 is a circuit diagram of a conventional switching power supply circuit which can be constructed on the basis of the invention previously proposed by the applicant of this application.
In the power supply circuit shown in FIG. 10, a full-wave rectifying circuit comprising a bridge rectifying circuit Di and a smoothing capacitor Ci is equipped as a rectifying and smoothing circuit to which commercial alternating power (alternating input voltage VAC) is input to achieve a DC input voltage, whereby the rectified and smoothed voltage Ei corresponding to the once level of the alternating input voltage VAC is achieved.
At the primary side of the power supply circuit, a self-excited type is constructed as a voltage resonance type converter circuit which carries out a single-end operation on the basis of a one-stone switching element Q1. In this case, a bipolar transistor having high resistance to voltage (BJT; junction type transistor) is used as the switching element Q1.
The base of the switching element Q1 is connected to the anode of the smoothing capacitor Ci (rectified and smoothed voltage Ei) through a starting resistor Rs to achieve base current at the starting time from a rectifying and smoothing line.
A drive winding NB which is provided at the primary side of an insulating converter transformer PIT so as to have a turn number of 1T (turn), and a series resonance circuit for self-excited resonance driving which comprises a series an inductor LB, a resonance capacitor CB and a base current limiting resistor RB is connected between the base of the switching element Q1 and the earth at the primary side. A switching frequency fs for turning on/off the switching element Q1 is generated by the self-excited circuit. For example, the switching frequency fs is set to 66 KHz by the series resonance circuit.
A route for clamp current flowing when the switching element Q1 is turned off is formed by a clamp diode DD1 inserted between the base of the switching element Q1 and the cathode (the earth at the primary side) of the smoothing capacitor Ci. The collector of the switching element Q1 is connected to one end of the primary winding N1 of the insulating converter transformer PIT, and the emitter thereof is grounded.
A parallel resonance capacitor Cr is connected between the collector and emitter of the switching element Q1 in parallel. In this case, a primary series resonance circuit of the voltage resonance type converter is formed by the capacitance of the parallel resonance capacitor Cr itself and the leakage inductance L1 of the primary winding N1 side of the insulating converter transformer PIT.
The insulating converter transformer PIT transmits the switching output of the switching element Q1 to the secondary side. The insulating converter transformer PIT is equipped with an EE-type core comprising two E-type cores of ferrite material or the like which are assembled such that both the magnetic legs thereof are confronted to each other, and the primary winding N1 and the secondary winding N2 are wound around the center magnetic leg of the EE-type core by using a divisional bobbin so as to be separated from each other. Further, the EE-type core is assembled so that a gap is formed in the center magnetic leg thereof, whereby loose coupling based on a required coupling coefficient is achieved.
One end of the primary winding N1 of the insulating converter transformer PIT is connected to the switching element Q1, and the other end thereof is connected to the anode of the smoothing capacitor Ci (rectified and smoothed voltage Ei). Accordingly, an alternating voltage having the period corresponding to the switching frequency occurs at the primary winding N1 when the switching output of the switching element Q1 is supplied to the primary winding Ni.
Further, at the secondary side of the insulating converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated at the secondary winding N2. In this case, a secondary parallel resonance capacitor C2 is connected to the secondary winding N2 in parallel, so that a parallel resonance circuit is formed by the leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary parallel resonance capacitor C2. The parallel resonance circuit sets the alternating voltage induced in the secondary winding N2 to a resonance voltage, so that a voltage resonance operation can be achieved at the secondary side. That is, the power supply circuit described above has the construction of a xe2x80x9ccomposite resonance type switching converterxe2x80x9d in which a parallel resonance circuit for setting the switching operation to a voltage resonance type is provided at the primary side and a parallel resonance circuit for achieving the voltage resonance operation is provided at the secondary side.
The secondary side of the power supply circuit thus constructed is equipped with a half-wave rectifying circuit comprising a secondary rectifying diode D01 and a smoothing capacitor C01 which are connected to the secondary winding N2, thereby achieving a main secondary DC output voltage E01 corresponding to substantially the once level as the alternating voltage induced in the secondary winding N2.
Further, in this case, an intermediate tap is provided to the secondary winding N2, and a half-wave rectifying circuit comprising a rectifying diode D02 and a smoothing capacitor C02 is connected to the winding between the tap output line of the secondary winding N2 and the earth at the secondary side as shown in FIG. 10 to generate and output a low secondary DC output voltage E02.
In the power supply circuit, an active clamp circuit is equipped to the secondary side. That is, an auxiliary switching element Q2 of MOS-FET, a clamp capacitor C3 and a clamp diode DD2 are equipped as the secondary active clamp circuit. Further, a drive winding Ng1, a capacitor Cg1 and a resistor Rg1 are equipped as a driving circuit system for driving the auxiliary switching element Q2.
A clamp diode DD2 is connected between the drain and source of the auxiliary switching element Q2 in parallel. As a connection manner, the anode of the clamp diode DD2 is connected to the source, and the cathode is connected to the drain.
The drain of the auxiliary switching element Q2 is connected to the connection point between the tap output line of the secondary winding N2 and the anode of the rectifying diode D02 through a clamp capacitor C3. Further, the source of the auxiliary switching element Q2 is connected to the earth at the secondary side.
Accordingly, the secondary active clamp circuit is constructed by connecting the clamp capacitor C3 to the parallel connection circuit comprising the auxiliary switching element Q2 and the clamp diode DD2 in series. The circuit thus formed is further connected to the secondary winding N2 in parallel.
As the driving circuit system of the auxiliary switching element Q2, a series connection circuit of a capacitor Cg1, a resistor Rg1 and a drive winding Ng1 is connected to the gate of the auxiliary switching element Q2 as shown in FIG. 10. The series connection circuit forms the self-excited driving circuit for the auxiliary switching element Q2. That is, a signal voltage VGS from the self-excited type driving circuit is applied to the gate of the switching element Q2 to perform the switching operation. In this case, the drive winding Ng1 is formed at the end portion side of the secondary winding N2, and the number of turns in this case is set to 1T (turn), for example.
Accordingly, the voltage induced by the alternating voltage achieved at the primary winding N1 occurs at the drive winding Ng1. Further, in this case, voltages achieved at the secondary winding N2 and the drive winding Ng1 are opposite in polarity because of the relationship of the winding direction between the windings N2 and Ng1.
In this embodiment, PWM control is carried out on the switching operation of the auxiliary switching element Q2 by a control circuit 1 equipped to the secondary side. That is, the secondary DC output voltage E01, E02 is supplied to the control circuit of an error amplifier, and the control circuit 1 applies the DC control voltage corresponding to E01, E02 to the gate of the auxiliary switching element Q2 to control the conduction angle of the auxiliary switching element Q2, whereby the DC output voltage is stabilized with respect to variations of the alternating input voltage VAC and the load power Po. For example, the main DC output voltage E01 is stabilized to 135V.
In the circuit construction described above, the insulating converter transformer PIT may be EE-40 under the condition that the gap is set to 1 mm, the number of turns of the primary winding N1 is set to 50T, the number of turns of the secondary winding N2 is set to 55T and the number of turns of the drive winding NB=Ng1 is set to 1T. Further, the resistance of the starting resistor Rs is set to 330 Kxcexa9, the resistance of the resistor RB is set to 1xcexa9, the capacitance of the capacitor CB is set to 0.33 xcexcF, the inductance of the inductor LB is set to 10 xcexcH, the capacitance of the resonance capacitor Cr is set to 0.012 xcexcF, the resistance of the resistor Rg1 is set to 10xcexa9, the capacitance of the resonance capacitor C2 is set to 0.012 xcexcF, the capacitance of the clamp capacitor C3 is set to 0.47 xcexcF, and the switching frequency fs is equal to 66 KHz.
FIG. 11 shows the operation waveforms of the respective parts at the alternating input voltage VAC of 100V and the load power Po of 200 W. FIG. 12 shows the operation waveforms of the respective parts under no load and at the alternating input voltage VAC of 100V. It is apparent from FIGS. 11 and 12 that the conduction angle (TON period) of the auxiliary switching element Q2 is controlled in accordance with the variation of the load power Po.
FIG. 13 shows the characteristic of the conduction angle (TON) and the AC/DC power conversion efficiency xcex7AC/DC with the variation of the load power Po, and FIG. 14 shows the characteristic of the conduction angle (TON) and the AC/DC power conversion efficiency xcex7AC/DC with the variation of the alternating input voltage VAC. As is apparent from FIGS. 13 and 14, the conduction angle (TON) of the auxiliary switching element Q2 with respect to the variation of the load power Po and the alternating input voltage VAC is controlled over the range from 4.6 xcexcs to 13 xcexcs.
As the conduction angle (TON) is increased due to reduction of the load power Po or increase of the alternating input voltage VAC, the conduction loss of the auxiliary switching element Q2 is increased and thus the AC/DC power conversion efficiency (xcex7AC/DC) is lowered.
Further, the primary side voltage resonance pulse voltage V1 occurring between both the ends of the main switching element Q1 is equal to 530V to 850V when the alternating input voltage VAC ranges from 90V to 144V, and thus the main switching element Q1 is required to have a voltage resistance to 900V. Further, the clamp voltage V2 at the secondary side is equal to 160V to 350V, and thus the auxiliary switching element Q2 is required to have a voltage resistance to 400V.
Current having a saw-tooth waveform flows in the switching elements Q1, Q2 under the no-load state as indicated by current IQ1, IC3 in FIG. 7. Therefore, these large amounts of current flow in the primary winding N1, the secondary winding N2 of the insulating converter transformer PIT and each switching element Q1, Q2, so that invalid current is increased and the input power at this time is equal to 13.5 W.
When stabilization is carried out on the main voltage (DC output main voltage) E01=135V with respect to the variations of the alternating input voltage VAC and the load current on the basis of the conduction angle control of the auxiliary switching element Q2 of the secondary active clamp circuit, the auxiliary DC output voltage (DC output low voltage) E02, which is equal to 15V for example, exhibits cross regulation of 15Vxc2x11.5V.
When the load current of the DC output low voltage E02 is varied by 0.5 A to 1.5 A, the cross regulation is further magnified to 15Vxc2x12.5V.
Therefore, in order to achieve a voltage-stabilized output voltage of 12V with a three-terminal regulator having a low saturation voltage, the power loss of (17.5xe2x88x9212.0)xc3x971.5=8.25 w at maximum occurs and thus the power conversion efficiency is further lowered.
As is understood from the foregoing description, the following problems (1) to (4) occur in the construction of the power supply circuit of FIG. 10 as described above, that is, in the construction in which the switching frequency of the auxiliary switching element Q2 of the active clamp circuit is fixed and the DC output voltage is stabilized by controlling the conduction angle.
(1) the conduction time of the secondary auxiliary switching element Q2 is increased due to reduction of the load power Po and increase of the alternating input voltage VAC, and thus the power loss is also increased, resulting in reduction in efficiency.
(2) The peak value of the voltage resonance pulse voltage VI occurring in the primary switching element Q1 increases in proportion to the alternating input voltage VAC, so that the switching element Q1 has a voltage resistance to 900V in the AC 100V system and a voltage resistance to 1800V in the AC 200V system. Therefore, an expensive switching element Q1 must be prepared.
(3) Under the no-load state, the invalid power is large, and the voltage resonance pulse voltage V1 is larger than under a load-applied state, so that a heat radiating plate is needed to the auxiliary switching element Q2.
(4) Since the cross regulation of the DC output lower voltage E02 taken out from the secondary side is large, a three-terminal series regulator is required to stabilize the DC output low voltage E02, and thus the power loss is further increased. Therefore, the power conversion efficiency is lowered.
Therefore, in consideration of the foregoing problems, according to the present invention, there is provided a switching power supply circuit comprising: switching means having a main switching element for conducting a switching operation on a DC input voltage and outputting a switching result; an insulating converter transformer that is designed so as to provide a coupling coefficient required for the loose coupling between a primary side and a secondary side, and transmits the output of the switching means achieved at the primary side to the secondary side; a primary parallel resonance circuit constructed by connecting a primary parallel resonance capacitor to a primary winding wound around the converter transformer equivalently and in parallel; a first secondary parallel resonance circuit constructed by connecting a first secondary parallel resonance capacitor to a secondary winding wound around the insulating converter transformer in parallel; first DC output voltage generating means for receiving an alternating voltage achieved at the secondary winding wound around the insulating converter transformer and conducting a rectifying operation on the alternating voltage thus received to a DC output main voltage; first stabilized voltage control means for performing voltage-stabilizing control on the DC output main voltage by controlling the switching frequency and conduction angle of the main switching element in accordance with the level of the DC output main voltage; a second secondary parallel resonance circuit constructed by connecting, in parallel, a second secondary parallel resonance capacitor to the winding portion between an intermediate tap of the secondary winding wound around the insulating converter transformer and the earth at the secondary side; second DC output voltage generating means for receiving an alternating voltage achieved at the intermediate tap of the secondary winding wound around the insulating converter transformer and conducting a rectifying operation on the alternating voltage thus received to achieve a DC output low voltage; active clamp means constructed by connecting, in parallel, a series connection circuit comprising a clamp capacitor and an auxiliary switching element to the winding portion between the intermediate tap of the secondary winding wound around the insulating converter transformer and the earth at the secondary side; and second voltage-stabilizing control means for executing the conduction angle control of the auxiliary switching element in accordance with the level of the DC output low voltage to perform the voltage-stabilizing control on the DC output low voltage.
In the switching power supply circuit thus constructed, there is achieved a so-called composite resonance type switching converter in which the primary resonance circuit to form the voltage resonance type converter is provided at the primary side, and the secondary resonance circuit comprising the secondary winding and the secondary parallel resonance capacitor is provided at the secondary side. On the basis of this construction, the active clamp means for stabilizing the DC output low voltage taken from the intermediate tap of the secondary winding is provided at the secondary side, and the conduction angle of the auxiliary main switching element is controlled in accordance with the level of the DC output low voltage, whereby the power loss due to the cross regulation of the DC output low voltage can be reduced and the power conversion efficiency can be enhanced.